With the explosive growth of the cellular phone industry, the need has arisen to reduce cost and power consumption of mobile handsets. To keep costs down, the entire radio, including memory, application processor, digital baseband processor, analog baseband and RF circuits, would ideally be all integrated onto a single silicon die with a minimal count of external components. The use of low-voltage deep submicron CMOS processes allows for an unprecedented degree of scaling and integration in digital circuitry, but complicates implementation of traditional RF circuits. Furthermore, any mask adders for RF/analog circuits are not acceptable from a fabrication cost standpoint.
Consequently, a strong incentive has arisen to find digital architectural solutions to the RF functions. Areas currently in focus are phase/frequency and amplitude modulations of an RF carrier realized using a digitally-controlled oscillator (DCO) and a digitally-controlled power amplifier (DPA) circuits, respectively. They are digitally-intensive equivalents of the conventional voltage-controlled oscillator (VCO) and power amplifier (PA) driver circuits. Due to the fine feature size and high switching speed of the modern CMOS technology, the respective digital-to-frequency conversion (DFC) and digital-to-RF-amplitude conversion (DRAC) transfer functions could be made very linear and of high dynamic range.
A block diagram illustrating an example prior art polar transmitter is shown in FIG. 1. The polar transmitter, generally referenced 10, comprises CORDIC and polar signal processing block 12, digital to frequency conversion block (DFC) 14 and Digital to RF amplitude conversion block (DRAC) 16. The DFC 14 comprises a modulator 22 and digitally controlled oscillator (DCO) 24. The DRAC 16 comprises a modulator 18 and digital power amplifier (DPA) 20.
The I and Q samples of the Cartesian coordinate system generated in a digital baseband (DBB) are converted through CORDIC algorithm 12 into amplitude and phase samples of the polar coordinate system. The phase is then differentiated to obtain frequency deviation. The polar signals are then conditioned through signal processing to sufficiently increase the sampling rate in order to reduce the quantization noise density and lessen the effects of the modulating spectrum replicas. The frequency deviation output signal is fed into the DCO based DFC 14, which produces a phase modulated (PM) digital carrieryPM(t)=sgn(cos(ω0t+θ[k]))   (1)where sgn(x)=1 for x≧0 and sgn(x)=−1 for x<0, ω0=2πf0 is the angular RF carrier frequency, and θ[k] is the modulating baseband phase of the kth sample. The phase θ(t)=∫−∞t f(t)dt is an integral of frequency deviation, where t=k·T0 with T0 being the sampling period.
The amplitude modulation (AM) signal controls the envelope of the phase-modulated carrier by means of the DPA based DRAC 16. Higher-order harmonics of the digital carrier are filtered out by a matching network so that the sgn( ) operator is dropped. The composite DPA output comprises the desired RF output spectrum.yRF(t)=a[k]·cos(ω0t+θ[k])   (2)where, a[k] is the modulating baseband amplitude of the kth sample.
While digital polar modulated transmitters have been demonstrated for GSM, GPRS, EDGE (GGE), their usage for 3G (WCDMA) and other wideband wireless standards remains a daunting task. Polar modulation relies on splitting the digital IQ baseband signal into a phase (i.e. frequency) and amplitude bit stream. The phase signal θ(or differentiated phase signal (f=Δθ/θt) ) is used to directly modulate a digitally controlled oscillator (DCO), the output of which is then combined with the amplitude signal ρ in a Digital Power Amplifier (DPA). The θ (or f=Δθ/Δt) component generated when passing the 3.96 MHz WCDMA IQ signal through a CORDIC spreads significantly due to the nonlinear (i.e. arctan) operation. The resulting signal is no longer band limited and thus theoretically infinite modulation of the oscillator is needed to represent this phase signal. Although, in a discrete time system such as this, the maximum frequency deviation will be limited to the sampling rate, it is still in the order of tens of MHz as shown in FIG. 2. Any truncation in phase data will degrade EVM. Tight modulation resolution has to be maintained in order to keep the frequency quantization noise much lower than electronic DCO phase noise.
Statistically, it can be shown that most of this signal can be represented in a bandwidth that is 100 times the signal bandwidth. This bandwidth, however, is too large a modulation range for a single oscillator (i.e. DCO) to handle while still providing the needed granularity (i.e. quantization step size, phase noise, etc.) and frequency coverage to span all frequency bands, including the typical bands of GSM-EU, GSM-US, PCS, DCS and IMT2K. Since the DCO modulation range is limited, phase data is truncated resulting in a severely degraded error vector magnitude (EVM).
This is further exacerbated by the fact that the DCO typically operates at 2× (for high frequency bands) or 4× (for low frequency bands) the actual desired output channel frequency. This implies that the DCO modulation range must be at least 4× the needed range. In practice, however, the modulation range must be even greater in order to compensate for coarse tuning step size, process, voltage and temperature (PVT) variations, etc.
Since the bandwidth requirements for existing GGE (i.e. 2G and 2.5G) polar transmitters are much smaller than that required for WCDMA and can thus be easily handled by the DCO. Therefore, a solution to the bandwidth problem described above, is to use multiple DCO circuits, one for each frequency band, corresponding to four DCO circuits. A disadvantage of this solution is that since the DCO circuit incorporates a large integrated circuit inductor, significant area would be consumed. Even if such a solution was constructed, it is not certain whether (1) the full modulation range (i.e. fine frequency step) could be achieved while keeping the DCO phase noise within specification or (2) whether the EVM would be degraded and compromised.
Another problem associated with highly integrated transceivers is injection pulling of the RF oscillator by the strong RF output if there is a small, but non-zero, frequency difference between the aggressor's (PPA or PA output signal) harmonic and the victim (RF oscillator's LC tank). The small frequency difference could be due, for example, to the AM modulation part of a polar transmitter, which performs both AM and PM/FM modulations. If, however, the aggressor harmonic frequency does not fall into the vicinity of the victim's critical frequency (e.g., resonating frequency of the LC tank), then the injection pulling will not be significant.
It is thus desirable to have a mechanism that overcomes the disadvantage of the prior art techniques. The mechanism should preferably be implementable as a simple, all digital implementation and be capable of enabling a polar transmitter to be used with wideband modulation schemes. More specifically, the mechanism should enable an oscillator having a limited bandwidth to be used with large modulation ranges required by wideband modulation schemes such as 3G WCDMA. In addition, in order to further avoid the DCO frequency pulling by the transmitter RF signal, the mechanism should preferably oscillate at a rational RF frequency multiplier (n/m, such as 4/3 or 3/4 of the RF frequency) so as to avoid frequency pulling.